In digital communication systems and digital broadcasting systems, a signal is transmitted as an electromagnetic waveform through a physical channel, such as air, to receivers. A channel effect may occur, which is not ideal. Undesired channel effects may include multipath reflection and propagation fading, leading to signal distortion. To address the channel fading phenomenon, a conventional approach has been proposed, which employs a transmit diversity technology. This approach includes, for example, multiple transmitting antennas, such that if a signal received from one transmitter antenna is in a deep fade, a stronger signal is able to be received from a different transmitter antenna in order to maintain communication. It is more practical to implement transmit diversity technology, by adding components at a base station than to implement receive diversity technology, which would require additional components at the remote receiving units, increasing their cost and size.
One example of the conventional approach is space-time block code (“STBC”), which has been discussed, for example, by Alamouti in “A Simple Transmit Diversity Technique for Wireless Communications,” IEEE Journal on Selected Areas in Communications, vol. 16, pp. 1451-1458 (October 1998). Transmit diversity has been a key technology adopted by modern communication systems in order to improve the system performance, such as data rate and reliability. Alamouti describes a simple transmit diversity scheme for improving the signal quality at the receiver side by simple processing across two transmit antennas on the transmitting side. The obtained diversity order is equal to applying maximal-ratio receiver combining (MRRC) with two antennas at the receiver. The scheme described by Alamouti may be generalized to two transmit antennas and M receive antennas to provide a diversity order of 2M.
For example, FIGS. 1 and 2 illustrate a space-time block coding technology according to Alamouti. FIG. 1 illustrates the baseband representation of the two-branch transmit diversity scheme 100 disclosed by Alamouti. The scheme incorporates two transmit antennas (102, 104) and one receive antenna (106). The scheme disclosed by Alamouti may be defined by three functions: (1) the encoding and transmission sequence of information symbols at the transmitter; (2) the combining scheme at the receiver; and (3) the decision rule for maximum likelihood detection.
Regarding the encoding and transmission sequence of information symbols at the transmitter function, at a given symbol period, two signals are simultaneously transmitted from the two antennas. The signal transmitted from antenna zero is denoted by s0 and from the antenna one by s1. During the next symbol period signal (−s1*) is transmitted from antenna zero, and signal s0* is transmitted from antenna one where * is the complex conjugate operation. As shown in Table I, the encoding is performed in space and time (space-time coding). The encoding, however, may also be performed in space and frequency. Instead of two adjacent symbol periods, two adjacent carriers may be used (space-frequency coding).
As shown in Table I, the channel at time t may be modeled by a complex multiplicative distortion h0(t) for the transmit antenna zero (102) and h1(t) for transmit antenna one (104).
TABLE Iantenna zeroantenna onetime ts0s1time t + T−s1*s0*
Assuming that fading is constant across two consecutive symbols, it can be formulated:h0(t)=h0(t+T)=h0=α0ejθ0 h1(t)=h1(t+T)=h1=α1ejθ1  (Eq. 1)where T is the symbol duration. The received signals can then be expressed as:r0=r(t)=h0s0+h1s1+n0 r1=r(t+T)=−h0s1*+h1s0*+n1  (Eq. 2)where r0 and r1 are the received signals at time t and t+T and n0 and n1 are complex random variables representing receiver noise and interference.
Regarding the combining scheme at the receiver function, the combiner 110 shown in FIG. 1 in communication with channel estimator 108, builds the following two combined signals that are input to the maximum likelihood detector 112:s0=h0*r0+h1r1*s1=h1*r0−h0r1*  (Eq. 3)Substituting (Eqs. 1 and 2) into Eq. 3, yields:s0=(α02+α12)s0+h0*n0+h1n1*s1=(α02+α12)s1h0n1*+h1*n0  (Eq. 4)
Regarding the maximum likelihood decision rule, the combined signals in Eq. 4 are then sent to the maximum likelihood detector 112, which, for each signals s0 and s1, uses a decision rule depending on a constellation which describes the mapping from information bits to complex symbols. The resulting combined signals in Eq. 4 are equivalent to that obtained from two branch MRRC (maximal-ratio receiver combining). The only difference between the resulting combined signals and the one obtained from two branch MRRC is phase rotations on the noise components, which do not degrade the effective signal to noise ration (SNR). Thus, the resulting diversity order from the new two-branch transmit diversity scheme with one receiver is equal to that of two-branch MRRC. Therefore, full diversity may be obtained by implementing Alamouti's STBC (space-time block code) approach. However, Alamouti's approach is based on the assumption of flat-fading channel and on the assumption that channel state information is known by receivers.
An approach discussed by Lee and Williams in “A Space-Time Transmitter Diversity Technique for Frequency Selective Fading Channels,” Proc. IEEE Sensor Array and Multichannel Signal Processing Workshop, Mar. 16-17, 2000, pp. 149-152 proposes a multiple-input multiple-output orthogonal frequency division multiplexing (MIMO-OFDM) scheme by combining Alamouti's space-time block coding (STBC) with multicarrier modulation. The scheme proposed by Lee and Williams may be used for both flat-fading and frequency selective fading channels.
Lee and Williams describe the simple transmitter diversity scheme proposed by Alamouti as adapted to an OFDM system, illustrating effectiveness of space-time OFDM, and achieving diversity gain over frequency selective fading channels. A block diagram 300 of the two-branch space-time OFDM transmitter diversity system disclosed by Lee and William is shown in FIG. 3.
FIG. 3 illustrates successive data symbol vectors at the output of the serial to parallel converter 302. These successive data symbol vectors may be considered one pair at a time. FIG. 3 illustrates the first vector in the pair as odd vector x0 and the second in the pair as the even vector x1. If x1 is the M-th block data symbol vector and x1 is the (M+1)-th block vector, they may be defined as:x0=[X(MN) . . . X(MN+N−1)]T x1=[X(MN+N) . . . X(MN+2N−1)]T  (Eq. 5)
At the first transmitter 304, x0 is transmitted through IFFT (Inverse Fast Fourier Transform) block 306 and cyclic prefix addition block 308, during the first time slot followed by −x1* in the second time slot. At the second transmitter, x1 is transmitted first through IFFT (Inverse Fast Fourier Transform) block 310 and cyclic prefix addition block 312 followed by x0*. The equivalent space-time block code transmission matrix may be expressed as:
      G    2    =      (                                        x            0                                                x            1                                                            -                          x              1              *                                                            x            0            *                                )  Thus, the entries of the transmission matrix are the OFDM symbol vectors, x0 and x1, and their conjugates.
Letting Λ0 and Λ1 (shown in FIG. 3 as outputs of channel estimator 324) be two diagonal matrices whose diagonal elements are the discrete Fourier transforms (DFTs) of the respective channel impulse responses, h0 and h1. Assuming that the channel responses are constant during the two time slots, the demodulated vectors in the corresponding time slots are determined as:Y0=Λ0X0+Λ1X1+Z0 Y1=−Λ0X1*+Λ1X0*+Z1  (Eq. 6)
Assuming the channel responses are known or can be estimated at the receiver 314, the decision variables are constructed by combining Y0, Y1 (shown in FIG. 3 as outputs of cyclic prefix removal block 316 and point Fast Fourier Transform (FFT) block 318), and the channel response matrices as{circumflex over (X)}0=Λ0*Y0+Λ1Y1*{circumflex over (X)}1=Λ1*Y0−Λ0Y1*  (Eq. 7)Substituting Eq. 6 into Eq. 7 yields:{circumflex over (X)}0=(|Λ0|2+|Λ1|2)X0+Λ0*Z0+Λ1Z1*{circumflex over (X)}1=(|Λ0|2+|Λ1|2)X1+Λ1*Z0+Λ0Z1*  (Eq. 8)Wherein {circumflex over (X)}[n] represents the output of the combiner and detector 320 and parallel to serial converter 322. These decision equations for the transmit diversity scheme 300 proposed by Lee and Williams are similar in form to that of a two-branch MRC receiver diversity system.
The transmit diversity technology described above was not adopted by any existing digital video/audio broadcasting systems, such as the Digital Video Broadcasting-Terrestrial (DVB-T), Digital Video Broadcasting-Handheld (DVB-H), and Integrated Services Digital Broadcasting-Terrestrial (ISDB-T) systems, because these standards were disseminated before the development of the transmit diversity technology. Thus, the performance of the DAB, ISDB-T, DVB-T, and DVB-H is limited, due to the single-transmit antenna structure scheme.
For a point-to-multipoint service in the application of digital video/audio broadcasting, transmit diversity technology achieves the best efficacy because all subscribers simultaneously enjoy improved performance at the cost of a small investment at the base station. However, present transmit diversity technologies (including the STBC schemes mentioned above) are not compatible with single transmit antenna structures. As a result, existing receivers may not be able to function properly if a transmit diversity technology is adopted by the transmitter of an existing broadcasting system.
Thus, there is a need for a scheme introducing the transmit diversity technology to existing digital video/audio broadcasting systems with backward compatibility to existing receivers.